Wireless terminal

ABSTRACT

A wireless terminal having antenna diversity comprises a transceiver coupled to a plurality of antenna feeds and a ground conductor ( 902 ), the antenna feeds being coupled directly to the ground conductor ( 902 ). In one embodiment the ground conductor is a conducting case ( 902 ). The coupling may be via parallel plate capacitors ( 504 ) formed by a respective plate ( 506 ) and a surface ( 908 ) of the case ( 902 ). The case ( 902 ) acts as an efficient, wideband radiator, eliminating the need for separate antennas. Slots ( 912 ) may be provided to increase the radiating bandwidth of the terminal and improve its diversity performance. Good diversity performance is obtained in a range of environments, whether the terminal is hand-held or free-standing.

FIELD OF THE INVENTION

The present invention relates to a wireless terminal providing antennadiversity, for example a mobile phone handset.

BACKGROUND OF THE INVENTION

Wireless terminals, such as mobile phone handsets, typically incorporateeither an external antenna, such as a normal mode helix or meander lineantenna, or an internal antenna, such as a Planar Inverted-F Antenna(PIFA) or similar.

Such antennas are small (relative to a wavelength) and therefore, owingto the fundamental limits of small antennas, narrowband. However,cellular radio communication systems typically have a fractionalbandwidth of 10% or more. To achieve such a bandwidth from a PIFA forexample requires a considerable volume, there being a directrelationship between the bandwidth of a patch antenna and its volume,but such a volume is not readily available with the current trendstowards small handsets. Hence, because of the limits referred to above,it is not feasible to achieve efficient wideband radiation from smallantennas in present-day wireless terminals.

A further problem with known antenna arrangements for wireless terminalsis that they are generally unbalanced, and therefore couple strongly tothe terminal case. As a result a significant amount of radiationemanates from the terminal itself rather than the antenna. A wirelessterminal in which an antenna feed is directly coupled to the terminalcase, thereby taking advantage of this situation, is disclosed in ourco-pending unpublished United Kingdom patent application 0108899.6(Applicant's reference PHGB010056). When fed appropriately, the terminalcase acts as an efficient, wideband radiator.

In many situations it is desirable for a wireless terminal to implementantenna diversity, whereby two or more antennas are used together toimprove performance over that which can be achieved with a singleantenna. In general, antenna diversity results in better reception,power savings and hence longer battery life. However, provision of twoor more conventional antennas in a wireless terminal, such as a mobilephone handset, requires a significant extra volume which is undesirablegiven the present trend to smaller and smaller handsets.

SUMMARY OF THE INVENTION

An object of the present invention is to provide a compact wirelessterminal having antenna diversity and efficient radiation propertiesover a wide bandwidth.

According to the present invention there is provided a wireless terminalcomprising a ground conductor and a transceiver coupled to a pluralityof antenna feeds, wherein each antenna feed is coupled directly to theground conductor.

Because the ground conductor (typically a handset body) is used as theradiating element, there is minimal extra volume required to implementantenna diversity (simply the volume occupied by a second capacitor orother coupling element). Hence, the present invention provides antennadiversity with a much-reduced volume requirement compared to knownarrangements, while also providing a significantly larger bandwidth.Although the use of two feeds to a common radiating element might beexpected to result in high correlation between the two antenna patterns,it is shown that in fact low correlation (and hence good diversityperformance) is achieved in practice.

The present invention is based upon the recognition, not present in theprior art, that the impedances of an antenna and a wireless handset aresimilar to those of an asymmetric dipole, which are separable, and onthe further recognition that the antenna impedance can be replaced witha non-radiating coupling element.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the present invention will now be described, by way ofexample, with reference to the accompanying drawings, wherein:

FIG. 1 shows a model of an asymmetrical dipole antenna, representing thecombination of an antenna and a wireless terminal;

FIG. 2 is a graph demonstrating the separability of the components ofthe impedance of an asymmetrical dipole;

FIG. 3 is an equivalent circuit of the combination of a handset and anantenna;

FIG. 4 is an equivalent circuit of a capacitively back-coupled handset;

FIG. 5 is a perspective view of a basic capacitively back-coupledhandset;

FIG. 6 is a graph of simulated return loss S₁₁ in dB against frequency fin MHz for the handset of FIG. 5;

FIG. 7 is a Smith chart showing the simulated impedance of the handsetof FIG. 5 over the frequency range 1000 to 2800 MHz;

FIG. 8 is a graph showing the simulated resistance of the handset ofFIG. 5;

FIG. 9 is a perspective view of a doubly-slotted capacitivelyback-coupled handset having two feeds;

FIG. 10 is a graph of simulated return loss S₁₁ in dB against frequencyf in MHz for one feed of the handset of FIG. 9;

FIG. 11 is a Smith chart showing the simulated impedance of one feed ofthe handset of FIG. 9 over the frequency range 1000 to 2800 MHz;

FIG. 12 is a graph of simulated return loss S₁₁ in dB against frequencyf in MHz for one feed of the handset of FIG. 9 with additional matching;

FIG. 13 is a Smith chart showing the simulated impedance of one feed ofthe handset of FIG. 9, with additional matching, over the frequencyrange 1000 to 2800 MHz;

FIG. 14 is a graph of simulated return loss S11 in dB against frequencyf in MHz for one feed of the handset of FIG. 9 with additional matchingand held in a hand; and

FIG. 15 is a Smith chart showing the simulated impedance of one feed ofthe handset of FIG. 9, with additional matching and held in a hand, overthe frequency range 1000 to 2800 MHz.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

In the drawings the same reference numerals have been used to indicatecorresponding features.

FIG. 1 shows a model of the impedance seen by a transceiver, in transmitmode, in a wireless handset at its antenna feed point. The impedance ismodelled as an asymmetrical dipole, where the first arm 102 representsthe impedance of the antenna and the second arm 104 the impedance of thehandset, both arms being driven by a source 106. As shown in the figure,the impedance of such an arrangement is substantially equivalent to thesum of the impedance of each arm 102, 104 driven separately against avirtual ground 108. The model could equally well be used for receptionby replacing the source 106 by an impedance representing that of thetransceiver, although this is rather more difficult to simulate.

The validity of this model was checked by simulations using thewell-known NEC (Numerical Electromagnetics Code) with the first arm 102having a length of 40 mm and a diameter of 1 mm and the second arm 104having a length of 80 mm and a diameter of 1 mm. FIG. 2 shows theresults for the real and imaginary parts of the impedance (R+jX) of thecombined arrangement (Ref R and Ref X) together with results obtained bysimulating the impedances separately and summing the result. It can beseen that the results of the simulations are quite close. The onlysignificant deviation is in the region of half-wave resonance, when theimpedance is difficult to simulate accurately.

An equivalent circuit for the combination of an antenna and a handset,as seen from the antenna feed point, is shown in FIG. 3. R₁ and jX₁represent the impedance of the antenna, while R₂ and jX₂ represent theimpedance of the handset. From this equivalent circuit it can be deducedthat the ratio of power radiated by the antenna, P₁, and the handset,P₂, is given by $\frac{P_{1}}{P_{2}} = \frac{R_{1}}{R_{2}}$

If the size of the antenna is reduced, its radiation resistance R₁ willalso reduce. If the antenna becomes infinitesimally small its radiationresistance R₁ will fall to zero and all of the radiation will come fromthe handset. This situation can be made beneficial if the handsetimpedance is suitable for the source 106 driving it and if thecapacitive reactance of the infinitesimal antenna can be minimised byincreasing the capacitive back-coupling to the handset.

With these modifications, the equivalent circuit is modified to thatshown in FIG. 4. The antenna has therefore been replaced with aphysically very small back-coupling capacitor, designed to have a largecapacitance for maximum coupling and minimum reactance. The residualreactance of the back-coupling capacitor can be tuned out with a simplematching circuit. By correct design of the handset, the resultingbandwidth can be much greater than with a conventional antenna andhandset combination, because the handset acts as a low Q radiatingelement (simulations show that a typical Q is around 1), whereasconventional antennas typically have a Q of around 50.

A basic embodiment of a capacitively back-coupled handset is shown inFIG. 5. A handset 502 has dimensions of 10×40×100 mm, typical of moderncellular handsets. A parallel plate capacitor 504, having dimensions2×10×10 mm, is formed by mounting a 10×10 mm plate 506 2 mm above thetop edge 508 of the handset 502, in the position normally occupied by amuch larger antenna. The resultant capacitance is about 0.5 pF,representing a compromise between capacitance (which would be increasedby reducing the separation of the handset 502 and plate 506) andcoupling effectiveness (which depends on the separation of the handset502 and plate 506). The capacitor is fed via a support 510, which isinsulated from the handset case 502.

The return loss S₁₁ of this embodiment after matching was simulatedusing the High Frequency Structure Simulator (HFSS), available fromAnsoft Corporation, with the results shown in FIG. 6 for frequencies fbetween 1000 and 2800 MHz. A conventional two inductor “L” network wasused to match at 1900 MHz. The resultant bandwidth at 7 dB return loss(corresponding to approximately 90% of input power radiated) isapproximately 60 MHz, or 3%, which is useful but not as large as wasrequired. A Smith chart illustrating the simulated impedance of thisembodiment over the same frequency range is shown in FIG. 7.

The low bandwidth is because the combination of the handset 502 andcapacitor 504 present an impedance of approximately 3-j90Ω at 1900 MHz.FIG. 8 shows the resistance variation, over the same frequency range asbefore, simulated using HFSS. This can be improved by redesigning thecase to increase the resistance, for example by the use of a slot or anarrower handset, as discussed in our co-pending unpublished UnitedKingdom patent application 0019335.9

In order to provide antenna diversity, at least two coupling elementsare required. An example of how this can be done is shown in FIG. 9. Adiversity handset 902 has a conducting case having dimensions of10×40×100 mm, into which two slots 912 have been cut. Each slot 912 hasa width of 3 mm and a depth of 29.5 mm and is placed 12 mm in from aside of the handset 902. As in the previous embodiment, a capacitor 504is formed from a plate 506, having dimensions 10×10 mm, mounted 4 mmabove the top surface 908 of the handset 902 on a support 510.

The return loss S₁₁ of this embodiment was simulated using HFSS, withthe results shown in FIG. 10 for frequencies f between 1000 and 2800MHz. In the simulation one capacitor 504 was fed directly, withoutmatching, while the other capacitor 504 was left open circuit. There aretwo resonances present, one centred at 1.83 GHz and the other at 2.24GHz. The first resonance is similar to that which would be achieved ifonly one capacitor 504 and slot 912 were present, as shown in ourco-pending unpublished UK patent application 0019335.9. The secondresonance is due to the presence of an additional slot 912. The centrefrequency of the first resonance is reduced by the presence of a secondslot 912, and hence the length of the slots 912 is reduced compared toan embodiment having a single slot. A Smith chart illustrating thesimulated impedance of this embodiment over the same frequency range isshown in FIG. 11. The rapid changes in impedance in the Smith chartreflect the narrow-band nature of the second resonance.

The response of this embodiment can be improved by matching. Simulationswere performed using a similar two inductor matching network to thatemployed in the basic embodiment, but matching both feedssimultaneously. This would be used in a dual receiver diversityarchitecture, where both antennas are available simultaneously. Similarperformance could be obtained with one feed connected and matched whilethe other is disconnected or loaded with another impedance, as would beused in a switched diversity configuration.

Results for the return loss S₁₁ are shown in FIG. 12 for frequencies fbetween 1000 and 2800 MHz. The resultant bandwidth at 7 dB return lossis now approximately 750 MHz, or nearly 40%. This is more than enough tocover UMTS and DCS 1800 bands simultaneously, which require coveragefrom 1710 to 2170 MHz. A Smith chart illustrating the simulatedimpedance of this embodiment over the same frequency range is shown inFIG. 13.

Further simulations were performed in which the handset was hand-held,with a 1 cm-thick hand placed around the lowest 60 mm of the handset andsurrounding it on three sides. The hand was simulated as a uniformvolume of complex dielectric material, having a dielectric constant of49 and a conductivity of 1.6 S/m at 1900 MHz. Results for return lossS₁₁ and a Smith chart are shown in FIGS. 14 and 15 respectively. Despitethe handset acting as part of the radiating system, the antennaefficiency is only reduced by 27% (computed as the ratio of input powerto power integrated over the problem space boundary in the simulation.)This is a similar reduction in efficiency to that found whenconventional handsets are hand-held.

For antenna diversity to be useful, it is necessary that the radiationpatterns of the individual antennas are sufficiently decorrelated. Acorrelation of less than 0.7 is generally taken to indicate gooddiversity performance. The correlation of the handset 902 was computed,for matched feeds, at three frequencies across the operating band andfor a variety of usage scenarios, with the following results:

Frequency (MHz) Environment 1711 1918 2170 rural 0.58 0.21 0.63 suburban0.46 0.10 0.51 urban macro/microcell 0.45 0.10 0.50 urban picocell 0.460.11 0.51 outdoors to indoors 0.34 0.04 0.37 indoors 0.35 0.05 0.39

The correlation was also computed for a hand-held handset, with the handcovering the lower 60 mm of three sides of the handset 902. Thefollowing results were obtained:

Frequency (MHz) Environment 1711 1918 2170 rural 0.21 0.04 0.45 suburban0.14 0.05 0.46 urban macro/microcell 0.18 0.06 0.45 urban picocell 0.100.00 0.38 outdoors to indoors 0.06 0.02 0.30 indoors 0.09 0.01 0.31

The above results clearly demonstrate that good diversity performance isobtained in a range of environments over a wide bandwidth. Results wouldbe expected to be similarly good for the case of one capacitor 504 fedwith the other capacitor 504 terminated in an unmatched load, as wouldbe the case for switched diversity.

The diversity embodiment described above made use of slots 912 in thehandset case 902 to enhance the feed match for coverage of both DCS1800and UMTS bands. Other embodiments are possible (including those withouthandset slots) which may trade off bandwidth against volume for example.When slots are provided, they may be extended to run the full length ofthe handset, and additional slots may also be provided for enhancedmulti-band operation. The function of the slots 912 in the diversityembodiment described above is to provide an impedance transformation sothat the antenna feed provides a reasonable match to 50Ω. Adequatediversity performance should be achieved providing that the antennafeeds are separated sufficiently on the ground conductor 902 (forexample those in FIG. 9 are separated by approximately 0.2 wavelengthsat 1711 MHz).

The embodiments disclosed above are based on capacitive coupling.However, any other sacrificial (non-radiating) coupling element could beused instead, for example inductive coupling. Also, the coupling elementcould be altered in order to aid impedance matching. For example,capacitive coupling could be achieved via an inductive element. Thiswould allow easier matching to yield a more wideband response.

In the above embodiments a conducting handset case has been theradiating element. However, other ground conductors in a wirelessterminal could perform a similar function. Examples include conductorsused for EMC shielding and an area of Printed Circuit Board (PCB)metallisation, for example a ground plane.

From reading the present disclosure, other modifications will beapparent to persons skilled in the art. Such modifications may involveother features which are already known in the design, manufacture anduse of wireless terminals and component parts thereof, and which may beused instead of or in addition to features already described herein.

In the present specification and claims the word “a” or “an” precedingan element does not exclude the presence of a plurality of suchelements. Further, the word “comprising” does not exclude the presenceof other elements or steps than those listed.

What is claimed is:
 1. A wireless terminal comprising a ground conductorand a plurality of antenna feeds, wherein each antenna feed is coupleddirectly to the ground conductor, wherein each side of the groundconductor has a surface area greater than the surface area of each ofthe plurality of antenna feeds, wherein the ground conductor has atleast one uni-directional slot extending parallel to the longitudinalaxis of the terminal, and wherein the slot provides a gap between theplurality of antenna feeds.
 2. The terminal as claimed in claim 1,characterised in that each antenna feed is coupled to the groundconductor via a capacitor.
 3. The terminal as claimed in claim 2,characterised in that the capacitor is a parallel plate capacitor formedby a conducting plate and a portion of the ground conductor.
 4. Theterminal as claimed in claim 1, characterised in that the at least oneuni-directional slot extending parallel to the longitudinal axisprovides a tuning fork configuration having at least three tines at atleast one end of the ground conductor.
 5. The terminal as claimed inclaim 1, characterised in that a first of the at least oneuni-directional slot extending parallel to the longitudinal axis isparallel to a second of the at least one uni-directional slot.
 6. Theterminal as claimed in claim 1, characterised in that the groundconductor is a handset case.
 7. The terminal as claimed in claim 1,characterised in that a matching network is provided for each antennafeed.